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mic2176-1/-2/-3 wide input voltage, synchronous buck controllers featuring adaptive on-time control hyper speed control? family hype mlf and 4-0800 ? fax + 1 ( r speed control, superswitcher ii and any capacitor are trademarks of micrel, inc. micro leadframe are registered trademarks of amkor technology, inc. micrel inc. ? 2180 fortune drive ? san jose, ca 95131 ? usa ? tel +1 ( 408 ) 94 408) 474-1000 ? http://www.micrel.com novem m9999-111710-a ber 2010 general description the micrel mic2176-1/-2/-3 is a family of constant-frequency, synchronous buck controllers featuring a unique digitally modified adaptive on-time control architecture. the mic2176 family operates over an input supply range of 4.5v to 75v and can be used to supply up to 15a of output current. the output voltage is adjustable down to 0.8v with a guaranteed accuracy of 1%, and the device operates at a constant switching frequency of 100khz, 200khz, and 300khz. micrel?s hyper speed control tm architecture allows for ultra- fast transient response while reducing the output capacitance and also makes (high v in )/(low v out ) operation possible. this digitally modified adaptive t on ripple control architecture combines the advantages of fixed-frequency operation and fast transient response in a single device. the mic2176 offers a full suite of protection features to ensure protection of the ic during fault conditions. these include undervoltage lockout to ensure proper operation under power-sag conditions, internal soft-start to reduce inrush current, fold-back current limit, ?hiccup? mode short- circuit protection and thermal shutdown. all support documentation can be found on micrel?s web site at: www.micrel.com . features ? hyper speed control tm architecture enables - high delta v operation (v in = 75v and v out = 1.2v) - small output capacitance ? 4.5v to 75v input voltage ? output down to 0.8v with 1% accuracy ? any capacitor tm stable - zero-esr to high-esr output capacitance ? 100khz/200khz/300khz switching frequency ? internal compensation ? 6ms internal soft-start ? foldback current limit and ?hiccup? mode short-circuit protection ? thermal shutdown ? supports safe start-up into a pre-biased output ? ?40 c to +125 c junction temperature range ? available in 10-pin msop package applications ? distributed power systems ? networking/telecom infrastructure ? printers, scanners, graphic cards and video cards ___________________________________________________________________________________________________________ typical application mic2176-2 adjustable output 200khz buck converter efficiency vs. output current 40 45 50 55 60 65 70 75 80 85 90 95 012345 output current (a) efficiency (%) 28v in 48v in 60v in mic2176-2 v out = 3.3v v dd = 5v linear
micrel, inc. mic2176 november 2010 2 m9999-111710-a ordering information part number output voltage switching frequency junction temperature range package lead finish MIC2176-1YMM adjustable 100khz ?40c to +125c 10-pin msop pb-free mic2176-2ymm adjustable 200khz ?40c to +125c 10-pin msop pb-free mic2176-3ymm adjustable 300khz ?40c to +125c 10-pin msop pb-free pin configuration 10-pin msop (mm) pin description pin number pin name pin function 1 hsd high-side mosfet drain connection (input): the hs d pin in the input of the adaptive on-time control circuitry. a 0.1uf ceramic capacitor betw een the hsd pin and the power ground (pgnd) is required and must be place as close as possible to the ic. 2 en enable (input): a logic level control of the outpu t. the en pin is cmos compatible. logic high or floating = enable, logic low = shut down. in the off state, the v dd supply current of the device is reduced (typically 0.7ma). do not connect the en pin to the hsd pin. 3 fb feedback (input): input to the tran sconductance amplifier of the cont rol loop. the fb pin is regulated to 0.8v. a resistor divider connecting the feedback to the output is used to adjust the desired output voltage. 4 gnd signal ground. gnd is the ground pa th for the device bias voltage v dd and the control circuitry. the loop for the signal ground should be separ ate from the power ground (pgnd) loop. 5 vdd v dd bias (input): power to the internal referenc e and control sections of the mic2176. the v dd operating voltage range is from 4.5v to 5.5v. a 1f ceramic capacitor from the vdd pin to the pgnd pin must be placed next to the ic. 6 dl low-side drive (output): high-current driver output for external low-side mosfet. the dl driving voltage swings from ground to v dd . 7 pgnd power ground. pgnd is the ground path for t he buck converter power stage. the pgnd pin connects to the sources of low-side n-channel ex ternal mosfets, the ne gative terminals of input capacitors, and the negative terminals of output capacitors. the loop for the power ground should be as small as possible and separate from the signal ground (gnd) loop. 8 dh high-side drive (output): high-curr ent driver output for external high-side mosfet. the dh driving voltage is floating on the switch node voltage (v sw ). adding a small resistor between dh pin and the gate of the high-side n-channel mosfets can slow down the turn-on and turn-off time of the mosfets.
micrel, inc. mic2176 november 2010 3 m9999-111710-a pin description (continued) pin number pin name pin function 9 sw switch node and current-sense input (input): high cu rrent output driver return. the sw pin connects directly to the switch node. due to the high-spee d switching on this pin, the sw pin should be routed away from sensitive nodes. the sw pin also senses the current by monitoring the voltage across the low-side mosfet during off time. in order to sense the current accurately, connect the low-side mosfet drain to the sw pin using a kelvin connection. 10 bst boost (output): bootstrappe d voltage to the high-side n-channel internal mosfet driver. a schottky diode is connected between the vdd pin an d the bst pin. a boost capacitor of 0.1 f is connected between the bst pin and the sw pin. adding a small resistor in series with the bst pin can slow down the turn-on time of high-side n-channel mosfets.
micrel, inc. mic2176 november 2010 4 m9999-111710-a absolute maximum ratings (1, 2) v hsd to pgnd................................................ ? 0.3v to +76v v dd to pgnd ................................................... ? 0.3v to +6v v sw to pgnd...................................... ? 0.3v to (v hsd +0.3v) v bst to v sw ........................................................ ? 0.3v to 6v v bst to pgnd .................................................. ? 0.3v to 82v v en to pgnd ...................................... ? 0.3v to (v dd + 0.3v) v fb to pgnd....................................... ? 0.3v to (v dd + 0.3v) pgnd to gnd .............................................. ? 0.3v to +0.3v junction temperature .............................................. +150c storage temperature (t s )......................... ? 65 c to +150 c lead temperature (solde ring, 10sec ) ........................ 260c operating ratings (3) supply voltage (v hsd ) ....................................... 4.5v to 75v bias voltage (v dd )............................................ 4.5v to 5.5v enable input (v en ) ................................................. 0v to v dd junction temperature (t j ) ........................ ? 40 c to +125 c junction thermal resistance msop ( ja ) ..................................................130. 5c/w continuous power dissipation (derate 5.6mw/c above 70c) (t a = 70c) ........................................................421mw esd (human body mode) .......................................... 1.5kv electrical characteristics (4) v in = v hsd = 48v, v dd = 5v; v bst ? v sw = 5v; t a = 25c, unless noted. bold values indicate ? 40c ? t j ? +125c. parameter condition min. typ. max. units power supply input hsd voltage range (v hsd ) (5) 4.5 75 v v dd bias voltage operating bias voltage (v dd ) 4.5 5 5.5 v undervoltage lockout trip level v dd rising 3.2 3.85 4.45 v uvlo hysteresis 370 mv quiescent supply current v fb = 1.5v 1.4 3 ma shutdown supply current sw = unconnected, v en = 0v 0.7 2 ma reference 0c ? t j ? 85c (1.0%) 0.792 0.8 0.808 feedback reference voltage -40c ? t j ? 125c (1.5%) 0.788 0.8 0.812 v fb bias current v fb = 0.8v 5 500 na enable control en logic level high 4.5v < v dd < 5.5v 1.2 0.85 v en logic level low 4.5v < v dd < 5.5v 0.78 0.4 v en bias current v en = 0v 50 100 a oscillator mic2176-1 75 100 125 mic2176-2 150 200 250 switching frequency (6) mic2176-3 225 300 375 khz mic2176-1, v fb = 0v, hsd=4v, v o = 3.3v 96 mic2176-2, v fb = 0v, hsd=4v, v o = 3.3v 93 maximum duty cycle (7) mic2176-3, v fb = 0v, hsd=4v, v o = 3.3v 89 % minimum duty cycle v fb > 0.8v 0 % minimum off-time 360 ns minimum on-time 60 ns
micrel, inc. mic2176 november 2010 5 m9999-111710-a electrical characteristics (4) (continued) v in = v hsd = 48v, v dd = 5v; v bst ? v sw = 5v; t a = 25c, unless noted. bold values indicate ? 40c ? t j ? +125c. parameter condition min. typ. max. units soft start soft-start time 6 ms short circuit protection current-limit threshold v fb = 0.8v 103 130 162 mv short-circuit threshold v fb = 0v 19 48 77 mv fet drivers dh, dl output low voltage i sink = 10ma 0.1 v dh, dl output high voltage i source = 10ma v dd - 0.1v or v bst - 0.1v v dh on-resistance, high state 2.1 3.3 ? dh on-resistance, low state 1.8 3.3 ? dl on-resistance, high state 1.8 3.3 ? dl on-resistance, low state 1.2 2.3 ? sw leakage current v sw = 48v, v dd = 5v, v bst = 53v 55 a hsd leakage current v sw = 48v, v dd = 5v, v bst = 53v 55 a thermal protection over-temperature shutdown t j rising 160 c over-temperature shutdown hysteresis 25 c notes: 1. exceeding the absolute maximum rating may damage the device. 2. devices are esd sensitive. handling pr ecautions recommended. human body model, 1.5k ? in series with 100pf. 3. the device is not guaranteed to function outside operating range. 4. specification for packaged product only. 5. the application is fully functional at low v dd (supply of the control section) if the external mosfets have enough low voltage v th . 6. measured in test mode. 7. the maximum duty-cycle is limited by the fixed mandatory off-time t off of typically 360ns.
micrel, inc. mic2176 november 2010 6 m9999-111710-a typical characteristics v in operating supply current vs. input voltage 0 5 10 15 20 25 30 0 10203040506070 input voltage (v) v in shutdown current vs. input voltage 0 5 10 15 20 25 30 35 40 0 10203040506070 input voltage (v) v dd operating supply current vs. input voltage 0 2 4 6 8 10 shutdown current (a) supply current (ma) supply current (ma) mic2176-2 v out = 3.3v i out = 0a v dd = 5v sw itching mic2176-2 v out = 3.3v i out = 0a v dd = 5v switching v dd = 5v v en = 0v 0 10203040506070 input voltage (v) feedback voltage vs. input voltage 0.792 0.794 0.796 0.798 0.800 0.802 0.804 0.806 0.808 0 10203040506070 input voltage (v) current limit vs. input voltage 0 5 10 15 20 current limit (a) total regulation vs. input voltage 0.0% 0.2% 0.4% 0.6% 0.8% 1.0% total regulation (%) feedback voltage (v) 0 10203040506070 input voltage (v) v out = 3.3v v dd = 5v i out = 0a to 5a v out = 3.3v v dd = 5v i out = 0a 0 10203040506070 input voltage (v) v out = 1.2v v dd = 5v switching frequency vs. input voltage 160 180 200 220 240 switching frequency (khz) 0 10203040506070 input voltage (v) mic2176-2 v out = 3.3v i out = 0a v dd = 5v
micrel, inc. mic2176 november 2010 7 m9999-111710-a typical characteristics (continued) v dd operating supply current vs. temperature 0 2 4 6 8 10 -50 -20 10 40 70 100 130 temperature (c) v dd shutdown current vs. temperature 0 0.2 0.4 0.6 0.8 1 -50 -20 10 40 70 100 130 temperature (c) v dd uvlo threshold vs. temperature 3.2 3.4 3.6 3.8 4.0 4.2 vdd threshold (v) rising supply current (ma) supply current (ma) mic2176-2 v in = 48v v out = 3.3v i out = 0a v dd = 5v switching v in = 48v i out = 0a v dd = 5v v en = 0v falling v in = 48v 3.0 -50 -20 10 40 70 100 130 temperature (c) v in shutdown current vs. temperature 0 5 10 15 20 25 30 -50 -20 10 40 70 100 130 temperature (c) current limit vs. temperature 0 5 10 15 20 v in operating supply current vs. temperature 10 12 14 16 18 20 22 -50 -20 10 40 70 100 130 temperature (c) supply current (ma) supply current (a) current limit (a) mic2176-2 v in = 48v v out = 3.3v i out = 0a v dd = 5v switching v in = 48v v dd = 5v i out = 0a v in = 48v v out = 3.3v v dd = 5v -50 -20 10 40 70 100 130 temperature (c) load regulation vs. temperature -0.2% 0.0% 0.2% 0.4% 0.6% 0.8% 1.0% -50 -20 10 40 70 100 130 temperature (c) feedback voltage vs. temperature 0.792 0.794 0.796 0.798 0.800 0.802 0.804 0.806 0.808 -50 -20 10 40 70 100 130 temperature (c) line regulation vs. temperature -0.2% 0.0% 0.2% 0.4% 0.6% 0.8% 1.0% line regulation (%) feeback voltage (v) -50 -20 10 40 70 100 130 temperature (c) v in = 6v to 70v v out = 3.3v i out = 0a v dd = 5v v in = 48v v out = 3.3v i out = 0a v dd = 5v load regulation (%) v in = 48v v out = 3.3v v dd = 5v i out = 0a to 5a switching frequency vs. temperature 160 180 200 220 240 switching frequency (khz) -50 -20 10 40 70 100 130 temperature (c) mic2176-1 v in = 48v v out = 3.3v i out = 0a v dd = 5v en bias current vs. temperature 0 20 40 60 80 100 en bias current (a) -50 -20 10 40 70 100 130 temperature (c) v in = 48v v en = 0v v dd = 5v
micrel, inc. mic2176 november 2010 8 m9999-111710-a typical characteristics (continued) efficiency vs. output current 40 45 50 55 60 65 70 75 80 85 90 95 012345 o u t p ut current ( a ) feedback voltage vs. output current 0.792 0.794 0.796 0.798 0.800 0.802 0.804 0.806 0.808 012345 output current (a) output voltage vs. output current 3.267 3.278 3.289 3.300 3.311 3.322 3.333 feedback voltage (v) v in = 48v v out = 3.3v v dd = 5v output voltage (v) 28v in efficiency (%) 48v in 60v in v in = 48v v out = 3.3v v dd = 5v mic2176-2 v out = 3.3v v dd = 5v linear 012345 output current (a) line regulation vs. output current -0.2% 0.0% 0.2% 0.4% 0.6% 0.8% 1.0% 012345 output current (a) switching frequency vs. output current 160 180 200 220 240 012345 output current (a) ic case temperature* (v in = 28v) vs. output current 0 20 40 60 case temperature (c) v in = 6v to 70v v out = 3.3v v dd = 5v switching frequency (khz) line regulation (%) mic2176-2 v in = 48v v out = 3.3v v dd = 5v 012345 output current (a) mic2176-2 v in = 28v v out = 3.3v v dd = 5v ic case temperature* (v in = 48v) vs. output current 0 20 40 60 80 case temperature (c) 012345 output current (a) mic2176-2 v in = 48v v out = 3.3v v dd = 5v ic case temperature* (v in = 60v) vs. output current 0 20 40 60 80 100 case temperature (c) 012345 output current (a) mic2176-2 v in = 60v v out = 3.3v v dd = 5v case temperature* : the temperature measurement was taken at the hottest point on the mic2176 case mounted on a 5 square inch pcb, see thermal measurement section. actual results will depend upon the size of the pcb, ambient temperature and proximity to other he at emitting components.
micrel, inc. mic2176 november 2010 9 m9999-111710-a typical characteristics (continued) efficiency (v in = 28v) vs. output current 60 65 70 75 80 85 90 95 efficiency (%) efficiency (v in = 60v) vs. output current 30 35 40 45 50 55 60 65 70 75 80 85 90 efficiency (%) 01234567 output current (a) 5.0v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v mic2176-2 vdd = 5v linea r efficiency (v in = 48v) vs. output current 40 45 50 55 60 65 70 75 80 85 90 efficiency (%) 01234567 output current (a) 5.0v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v mic2176-2 vdd = 5v linea r 01234567 output current (a) 5.0v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v mic2176-2 vdd = 5v linea r
micrel, inc. mic2176 november 2010 10 m9999-111710-a functional characteristics
micrel, inc. mic2176 november 2010 11 m9999-111710-a functional characteristics (continued)
micrel, inc. mic2176 november 2010 12 m9999-111710-a functional characteristics (continued)
micrel, inc. mic2176 november 2010 13 m9999-111710-a functional diagram figure 1. mic2176 functional diagram
micrel, inc. mic2176 november 2010 14 m9999-111710-a functional description the mic2176 is an adaptive on-time synchronous buck controller family built for high-input voltage and low output voltage applications. it is designed to operate over a wide input voltage range from, 4.5v to 75v and the output is adjustable with an external resistive divider. a digitally modified adaptive on-time control scheme is employed in to obtain a constant switching frequency and to simplify the control compensation. over-current protection is implemented by sensing low-side mosfet?s r ds(on) . the device features internal soft- start, enable, uvlo, and thermal shutdown. theory of operation figure 1 illustrates the block diagram of the mic2176. the output voltage is sensed by the mic2176 feedback pin fb via the voltage divider r1 and r2, and compared to a 0.8v reference voltage v ref at the error comparator through a low-gain transconductance (g m ) amplifier. if the feedback voltage decreases and the amplifier output is below 0.8v, then the error comparator will trigger the control logic and generate an on-time period. the on- time period length is predetermined by the ?fixed t on estimator? circuitry: swin out ed) on(estimat f v v = t (1) where v out is the output voltage, v in is the power stage input voltage, and f sw is the switching frequency (100khz for mic2176-1, 200 khz for mic2176-2, and 300khz for mic2176-3). at the end of the on-time period, the internal high-side driver turns off the high-side mosfet and the low-side driver turns on the low-side mosfet. the off-time period length depends upon the feedback voltage in most cases. when the feedback voltage decreases and the output of the g m amplifier is below 0.8v, the on-time period is triggered and the off-time period ends. if the off-time period determined by the feedback voltage is less than the minimum off-time t off(min) , which is about 360ns, the mic2176 control logic will apply the t off(min) instead. t off(min) is required to maintain enough energy in the boost capacitor (c bst ) to drive the high-side mosfet. the maximum duty cycle is obtained from the 360ns t off(min) : s s off(min) s max t 360ns 1 t tt d ?= ? = (2) where t s = 1/f sw . it is not recommended to use mic2176 with a off-time close to t off(min) during steady-state operation. the adpative on-time control scheme results in a constant switching frequency in the mic2176. the actual on-time and resulti ng switching frequency will vary with the different rising and falling times of the external mosfets. also, the minimum t on results in a lower switching frequency in high v in to v out applications, such as 48v to 1.0v. the minimum t on measured on the mic2176 evaluation board is about 60ns. during load transients, the switching frequency is changed due to the varying off-time. to illustrate the control lo op operation, we will analyze both the steady-state and load transient scenarios. for easy analysis, the gain of the g m amplifier is assumed to be 1. with this assumption, the inverting input of the error comparator is the same as the feedback voltage. figure 2 shows the mic2176 control loop timing during steady-state operation. during steady-state, the g m amplifier senses the feedback voltage ripple, which is proportional to the output voltage ripple plus injected voltage ripple, to trigger the on-time period. the on- time is predetermined by the t on estimator. the termination of the off-time is controlled by the feedback voltage. at the valley of the feedback voltage ripple, which occurs when v fb falls below v ref , the off period ends and the next on-time period is triggered through the control logic circuitry. figure 2. mic2176 control loop timing
micrel, inc. mic2176 november 2010 15 m9999-111710-a figure 3 shows the operati on of the mic2176 during a load transient. the output voltage drops due to the sudden load increase, which causes the v fb to be less than v ref . this will cause the error comparator to trigger an on-time period. at the end of the on-time period, a minimum off-time t off(min) is generated to charge c bst since the feedback voltage is still below v ref . then, the next on-time period is triggered due to the low feedback voltage. therefore, the sw itching frequency changes during the load transient, but returns to the nominal fixed frequency once the output has stabilized at the new load current level. with the va rying duty cycle and switching frequency, the output recovery time is fast and the output voltage deviation is small in mic2176 converter. figure 3. mic2176 load transient response unlike true current-mode control, the mic2176 uses the output voltage ripple to trigger an on-time period. the output voltage ripple is proportional to the inductor current ripple if the esr of the output capacitor is large enough. the mic2176 control loop has the advantage of eliminating the need for slope compensation. in order to meet the stabili ty requirements, the mic2176 feedback voltage ripple should be in phase with the inductor current ripple and large enough to be sensed by the g m amplifier and the error comparator. the recommended feedback voltage ripple is 20mv~100mv. if a low esr output capacitor is selected, then the feedback voltage ripple may be too small to be sensed by the g m amplifier and the error comparator. also, the output voltage ripple and t he feedback voltage ripple are not necessarily in phase with the inductor current ripple if the esr of the output capacitor is very low. in these cases, ripple injection is required to ensure proper operation. please refer to ?ripple injection? subsection in application information for more details about the ripple injection technique. soft-start soft-start reduces the power supply input surge current at startup by cont rolling the output voltage rise time. the input surge appears while the output capacitor is charged up. a slower output ri se time will draw a lower input surge current. the mic2176 implements an internal digital soft-start by making the 0.8v reference voltage v ref ramp from 0 to 100% in about 6ms with 9.7mv steps. therefore, the output voltage is controlled to increase slowly by a stair- case v fb ramp. once the soft-start cycle ends, the related circuitry is disabled to reduce current consumption. v dd must be powered up at the same time or after v in to make the soft-start function correctly. current limit the mic2176 uses the r ds(on) of the low-side power mosfet to sense over-current conditions. this method will avoid adding cost, boar d space and power losses taken by discrete current sense resistors. the low-side mosfet is used because it displays much lower parasitic oscillations during switching than the high-side mosfet. in each switching cycle of the mic2176 converter, the inductor current is sensed by monitoring the low-side mosfet in the off period. the sensed voltage is compared with a current-limit threshold voltage v cl after a blanking time of 150ns. if the sensed voltage is over v cl , which is 133mv typical at 0.8v v fb , then the mic2176 turns off the high-side and low-side mosfets and a soft-start sequence is triggered. this mode of operation is called ?hiccup mode? and its purpose is to protect the downstream load in case of a hard short. the current limit threshold v cl has a foldback characteristic related to the fb voltage. please refer to the ?typical characteristics? for the curve of current limit threshold vs. fb voltage percentage. the circuit in figure 4 illustrates the mic2176 current limiting circuit. figure 4. mic2176 curre nt limiting circuit
micrel, inc. mic2176 november 2010 16 m9999-111710-a using the typical v cl value of 130mv, the current-limit value is roughly estimated as: mosfet gate drive ds(on) cl r 130mv i (3) for designs where the current ripple is significant compared to the load current i out , or for low duty cycle operation, calculating the current limit i cl should take into account that one is sensing the peak inductor current and that there is a blanking delay of approximately 150ns. 2 ?i l t r l(pp) dly ds(on) ? v 130mv i out cl + = (4) l f d)(1v ?i sw out l(pp) ? = (5) the mic2176 high-side drive circuit is designed to switch an n-channel mosfet. figure 1 shows a bootstrap circuit, consisting of d1 (a schottky diode is recommended) and c bst . this circuit supplies energy to the high-side drive circuit. capacitor c bst is charged while the low-side mosfet is on and the voltage on the sw pin is approximately 0v. when the high-side mosfet driver is turned on, energy from c bst is used to turn the mosfet on. as the high-side mosfet turns on, the voltage on the sw pin increases to approximately v in . diode d1 is reverse biased and c bst floats high while continuing to keep the high-side mosfet on. the bias current of the high-side driver is less than 10ma so a 0.1 f to 1 f is sufficient to hold the gate voltage with minimal droop for the power stroke (high-side switching) cycle, i.e. ? bst = 10ma x 3.33 s/0.1 f = 333mv. when the low-side mosfet is turned back on, c bst is recharged through d1. a small resistor r g , which is in series with c bst , can be used to slow down the turn-on time of the high-side n-channel mosfet. where the drive voltage is derived from the v dd supply voltage. the nominal low-side gate drive voltage is v dd and the nominal high-side gate drive voltage is approximately v dd ? v diode , where v diode is the voltage drop across d1. an approximate 30ns delay between the high-side and low-side driver transitions is used to prevent current from simultaneously flowing unimpeded through both mosfets. v out = the output voltage t dly = current-limit blanking time, 150ns typical ? i l(pp) = inductor current ripple peak-to-peak value d = duty cycle f sw = switching frequency the mosfet r ds(on) varies 30% to 40% with temperature; therefore, it is recommended to add a 50% margin to i cl in the above equation to avoid false current limiting due to increased mosfet junction temperature rise. it is also recommended to connect sw pin directly to the drain of the low-side mosfet to accurately sense the mosfets r ds(on) .
micrel, inc. mic2176 november 2010 17 m9999-111710-a application information mosfet selection swg side]-g[high f q(avg) i = sw gs iss side]-g[low f v c (avg) i the mic2176 controller works from power stage input voltages of 4.5v to 73v and has an external 4.5v to 5.5v v in to provide power to turn the external n-channel power mosfets for the high- and low-side switches. for applications where v dd < 5v, it is necessary that the power mosfets used are sub-logic level and are in full conduction mode for v gs of 2.5v. for applications when v dd > 5v; logic-level mosfets, whose operation is specified at v gs = 4.5v must be used. there are different criteria for choosing the high-side and low-side mosfets. these differences are more significant at lower duty cycles. in such an application, the high-side mosfet is required to switch as quickly as possible to minimize transition losses, whereas the low-side mosfet can switch slower, but must handle larger rms currents. when the duty cycle approaches 50%, the current ca rrying capability of the high-side mosfet starts to become critical. it is important to note that the on-resistance of a mosfet increases with increasing temperature. a 75c rise in junction temperature will increase the channel resistance of the mosfet by 50% to 75% of the resistance specified at 25c. this change in resistance must be accounted for when calculating mosfet power dissipation and in calculating the value of current limit. total gate charge is the charge required to turn the mosfet on and off under specified operating conditions (v ds and v gs ). the gate charge is supplied by the mic2176 gate-drive circuit. at 300khz switching frequency, the gate charge can be a significant source of power dissipation in the mic2176. at low output load, this power dissipation is noticeable as a reduction in efficiency. the average current required to drive the high-side mosfet is: (6) where: i g[high-side] (avg) = average high-side mosfet gate current q g = total gate charge for the high-side mosfet taken from the manufacturer?s data sheet for v gs = v dd . f sw = switching frequency the low-side mosfet is turned on and off at v ds = 0 because an internal body diode or external freewheeling diode is conducting during this time. the switching loss for the low-side mosfet is usually negligible. also, the gate-drive current for the low-side mosfet is more accurately calculated using c iss at v ds = 0 instead of gate charge. for the low-side mosfet: (7) = since the current from the gate drive comes from the v dd , the power dissipated in the mic2176 due to gate drive is: (8) (avg)) i (avg) (i v p side]-g[low side] g[high- dd gatedrive + = ac conduction sw p p p + a convenient figure of merit for switching mosfets is the on resistance times the total gate charge r ds(on) q g . lower numbers translate into higher efficiency. low gate-charge logic-level mosfets are a good choice for use with the mic2176. also, the r ds(on) of the low-side mosfet will determine the current-limit value. please refer to ?current limit? subsection is functional description for more details. p arameters that are important to mosfet switch selection are: ? voltage rating ? on-resistance ? total gate charge the voltage ratings for the high-side and low-side mosfets are essentially equal to the power stage input voltage v hsd . a safety factor of 20% should be added to the v ds (max) of the mosfets to account for voltage spikes due to circuit parasitic elements. the power dissipated in the mosfets is the sum of the conduction losses during the on-time (p conduction ) and the switching losses during the period of time when the mosfets turn on and off (p ac ). (9) = ds(on) 2 sw(rms) conduction r i p = ac(on) ) ac(off ac p p p += (10) (11) where: r ds(on) = on-resistance of the mosfet switch d = duty cycle = v out / v hsd
micrel, inc. mic2176 november 2010 18 m9999-111710-a making the assumption that the turn-on and turn-off transition times are equal; the transition times can be approximated by: g hsd oss in iss t i vcvc t + = swtpkd hsd ac f t i) v(v p += (12) where: c iss and c oss are measured at v ds = 0 i g = gate-drive current the total high-side mosfet switching loss is: (13) where: t t = switching transition time v d = body diode drop (0.5v) f sw = switching frequency the high-side mosfet switching losses increase with the switching frequency and the input voltage v hsd . the low-side mosfet switching losses are negligible and can be ignored for these calculations. inductor selection values for inductance, peak, and rms currents are required to select the output inductor. the input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. generally, higher inductance values are used with higher input voltages. larger peak-to-peak ripple currents will increase the power dissipation in the inductor and mosfets. larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. a good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. the inductance value is calculated by equation 14: out(max) sw in(max) out in(max) out i20% f v )v (vv l ? = (14) where: f sw = switching frequency, 300khz 20% = ratio of ac ripple current to dc output current v in(max) = maximum power stage input voltage the peak-to-peak inductor current ripple is: l f v )v (vv i sw in(max) out in(max) out l(pp) ? = (15) the peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor current ripple. i l(pk) =i out(max) + 0.5 ? i l(pp) (16) the rms inductor current is used to calculate the i 2 r losses in the inductor. 12 ?i ii 2 l(pp) 2 out(max) l(rms) + = (17) maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. the high frequency operation of the mic2176 requires the use of ferrite materials for all but the most cost sensitive applications. lower cost iron powder cores may be used but the increase in core loss will reduce the efficiency of the power supply. this is especially noticeable at low output power. the winding resistance decreases efficiency at the higher output current levels. the winding resistance must be minimized although this usually comes at the expense of a larger inductor. the power dissipated in the inductor is equal to the sum of the core and copper losses. at higher output loads, the core losses are usually insignificant and can be ignored. at lower output currents, the core losses can be a significant contributor. core loss information is usually available from the magnetics vendor.
micrel, inc. mic2176 november 2010 19 m9999-111710-a copper loss in the inductor is calculated by equation 18: p inductor(cu) = i l(rms) 2 r winding (18) the resistance of the copper wire, r winding , increases with the temperature. the value of the winding resistance used should be at the operating temperature. p winding(ht) = r winding(20c) (1 + 0.0042 (t h ? t 20c )) (19) where: t h = temperature of wire under full load t 20c = ambient temperature r winding(20c) = room temperature winding resistance (usually specified by the manufacturer) output capacitor selection the type of the output capacitor is usually determined by its esr (equivalent series resistance). voltage and rms current capability are two other important factors for selecting the output capacitor. recommended capacitor types are tantalum, low-esr aluminum electrolytic, os- con and poscap. the output capacitor?s esr is usually the main cause of the output ripple. the output capacitor esr also affects the control loop from a stability point of view. the maximum value of esr is calculated: l(pp) out(pp) c ?i ?v esr out (20) where: v out(pp) = peak-to-peak output voltage ripple ? i l(pp) = peak-to-peak inductor current ripple the total output ripple is a combination of the esr and output capacitance. the total ripple is calculated in equation 21: () 2 c l(pp) 2 sw out l(pp) out(pp) out esr ?i 8fc ?i ?v + ? ? ? ? ? ? ? ? = (21) where: d = duty cycle c out = output capacitance value f sw = switching frequency as described in the ?theory of operation? subsection in functional description , the mic2176 requires at least 20mv peak-to-peak ripple at the fb pin to make the g m amplifier and the error comparator behave properly. also, the output voltage ripple should be in phase with the inductor current. therefore, the output voltage ripple caused by the output capacitors value should be much smaller than the ripple caused by the output capacitor esr. if low esr capacitors, such as ceramic capacitors, are selected as the output capacitors, a ripple injection method should be applied to provide the enough feedback voltage ripple. please refer to the ?ripple injection? subsecti on for more details. the voltage rating of the capacitor should be twice the output voltage for a tantalum and 20% greater for aluminum electrolytic or os-con. the output capacitor rms current is calculated in equation 22: 12 ?i i l(pp) (rms) c out = out out out c 2 (rms) c) diss(c esr i p = (22) the power dissipated in the output capacitor is: (23) input capacitor selection the input capacitor for the power stage input v in should be selected for ripple current rating and voltage rating. tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. a tantalum input capacitor?s voltage rating should be at least two times the maximum input voltage to maximize reliability. aluminum electrolytic, os-con, and multilayer polymer film capacitors can handle the higher inrush currents without voltage de-rating. the input voltage ripple will primarily depend on the input capacitor?s esr. the peak input current is equal to the peak inductor current, so: ? v in = i l(pk) esr cin (24)
micrel, inc. mic2176 november 2010 20 m9999-111710-a the input capacitor must be rated for the input current ripple. the rms value of input capacitor current is determined at the maximum output current. assuming the peak-to-peak inductor current ripple is low: (25) d)(1d ii out(max) cin(rms) ? the power dissipated in the input capacitor is: p diss(cin) = i cin(rms) 2 esr cin (26) voltage setting components the mic2176 requires two resistors to set the output voltage as shown in figure 5: figure 5. voltage-divider configuration the output voltage is determined by the equation: ) r2 r1 (1vv fb out += (27) where, v fb = 0.8v. a typical value of r1 can be between 3k and 10k . if r1 is too large, it may allow noise to be introduced into the voltage feedback loop. if r1 is too small in value, it will decrease the efficiency of the power supply, especially at light loads. once r1 is selected, r2 can be calculated using: fb out fb vv r1v r2 ? = (28) ripple injection the v fb ripple required for proper operation of the mic2176 g m amplifier and error comparator is 20mv to 100mv. however, the output voltage ripple is generally designed as 1% to 2% of the output voltage. for a low output voltage, such as a 1v, the output voltage ripple is only 10mv to 20mv, and the feedback voltage ripple is less than 20mv. if the feedback voltage ripple is so small that the g m amplifier and error comparator can?t sense it, then the mic2176 will lose control and the output voltage is not regulated. in order to have some amount of v fb ripple, a ripple injection method is applied for low output voltage ripple applications. th e applications are divided into three situations according to the amount of the feedback voltage ripple: 1. enough ripple at the feedback voltage due to the large esr of the output capacitors. as shown in figure 6a, the converter is stable without any ripple injection. the feedback voltage ripple is: (pp) l c fb(pp) ?i esr r2r1 r2 ?v out + = (pp) l fb(pp) ? iesr ?v (29) wh ere ? i l(pp) is the peak-to-peak value of the inductor current ripple. 2. inadequate ripple at the feedback voltage due to the small esr of the output capacitors. the output voltage ripple is fed into the fb pin through a feedforward capacitor c ff in this situation, as shown in figure 6b. the typical c ff value is between 1nf and 100nf. with the feedforward capacitor, the feedback voltage ripple is very close to the output voltage ripple: (30) 3. virtually no ripple at the fb pin voltage due to the very-low esr of the output capacitors: figure 6a. enough ripple at fb
micrel, inc. mic2176 november 2010 21 m9999-111710-a if the voltage divider resistors r1 and r2 are in the k ? range, a c ff of 1nf to 100nf can easily satisfy the large time constant requirements. also, a 100nf injection capacitor c inj is used in order to be considered as short for a wide range of the frequencies. the process of sizing the ripple injection resistor and capacitors is: step 1. select c ff to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. typical choice of c ff is 1nf to 100nf if r1 and r2 are in k ? range. figure 6b. inadequate ripple at fb step 2. select r inj according to the expected feedback voltage ripple using equation 24: figure 6c. invisible ripple at fb in this situation, the output voltage ripple is less than 20mv. therefore, additional ripple is injected into the fb pin from the switching node sw via a resistor r inj and a capacitor c inj , as shown in figure 6c. the injected ripple is: w u uuuu sw divin fb(pp) f 1 d)-(1dkv v (31) r1//r2 r r1//r2 k inj div (32) where: v in = power stage input voltage d = duty cycle f sw = switching frequency = (r1//r2//r inj ) u c ff in equations 21 and 22, it is assumed that the time constant associated with c ff must be much greater than the switching period: 1 t f 1 sw u ww (33) d)(1d f v v k sw in fb(pp) div u u u w (34) then the value of r inj is obtained as: 1) k 1 ((r1//r2) r div inj u (35) step 3. select c inj as 100nf, which could be considered as short for a wide range of the frequencies.
micrel, inc. mic2176 november 2010 22 m9999-111710-a pcb layout guidelines warning!!! to minimize emi and output noise, follow these layout recommendations. pcb layout is critical to achieve reliable, stable and efficient performance. a ground plane is required to control emi and minimize the inductance in power, signal and return paths. the following guidelines should be followed to insure proper operation of the mic2176 converter. ic ? the 1f ceramic capacitor, which is connected to the vdd pin, must be located right at the ic. the vdd pin is very noise sens itive and placement of the capacitor is very critical. use wide traces to connect to the vdd and pgnd pins. ? the signal ground pin (gnd) must be connected directly to the ground planes. do not route the gnd pin to the pgnd pin on the top layer. ? place the ic close to the point of load (pol). ? use fat traces to route the input and output power lines. ? signal and power grounds should be kept separate and connected at only one location. i nput capacitor ? place the input capacitor next. ? place the input capacitors on the same side of the board and as close to the mosfets as possible. ? place several vias to the ground plane close to the input capacitor ground terminal. ? use either x7r or x5r dielectric input capacitors. do not use y5v or z5u type capacitors. ? do not replace the ceramic input capacitor with any other type of capacitor. any type of capacitor can be placed in parallel with the input capacitor. ? if a tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. ? in ?hot-plug? applications, a tantalum or electrolytic bypass capacitor must be used to limit the over- voltage spike seen on the input supply with power is suddenly applied. r c snubber ? place the rc snubber on the same side of the board and as close to the sw pin as possible. inductor ? keep the inductor connection to the switch node (sw) short. ? do not route any digital lines underneath or close to the inductor. ? keep the switch node (sw) away from the feedback (fb) pin. ? the sw pin should be connected directly to the drain of the low-side mosfet to accurate sense the voltage across the low-side mosfet. ? to minimize noise, place a ground plane underneath the inductor. o utput capacitor ? use a wide trace to connect the output capacitor ground terminal to the input capacitor ground terminal. ? phase margin will change as the output capacitor value and esr changes. contact the factory if the output capacitor is different from what is shown in the bom. ? the feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. sensing a long high-current load trace can degrade the dc load regulation. mosfets ? low-side mosfet gate drive trace (dl pin to mosfet gate pin) must be short and routed over a ground plane. the ground plane should be the connection between the mosfet source and pgnd. ? chose a low-side mosfet with a high c gs /c gd ratio and a low internal gate resistance to minimize the effect of dv/dt inducted turn-on. ? do not put a resistor between the low-side mosfet gate drive output and the gate. ? use a 4.5v v gs rated mosfet. its higher gate threshold voltage is more immune to glitches than a 2.5v or 3.3v rated mosfet. mosfets that are rated for operation at less than 4.5v v gs should not be used.
micrel, inc. mic2176 november 2010 23 m9999-111710-a evaluation board schematic figure 7. schematic of mic2176 evaluation board (j1, j9, j12, r12, and r 13 are for testing purposes)
micrel, inc. mic2176 november 2010 24 m9999-111710-a bill of materials item part number manufacturer description qty c1 b41125a9336m epcos (1) 33f aluminum capacitor, smd, 100v 1 c2, c3 grm32er72a225k murata (2) 2.2f/100v ceramic capacitor, x7r, size 1210 2 c4 6sepc470m sanyo (3) 470f/6.3v oscon capacitor 1 c5, c15 grm32er60j104ka94d murata (2) 100f/6.3v ceramic capacitor, x7r, size 1210 2 c6, c7, c16 grm188r71h104ka94l murata (2) 0.1f/6.3v ceramic capacitor, x7r, size 0603 3 c8 grm188r70j105ka01d murata (2) 1f/6.3v ceramic capacitor, x7r, size 0603 1 c9, c10 grm188r72a104ka35d murata (2) 0.1f/100v ceramic capacitor, x7r, size 0603 2 c11 grm188r72a102ka01d murata (2) 1nf/100v cermiac capacitor, x7r, size 0603 1 c12 grm188r71h103k murata (2) 10nf/50v ceramic capacitor, x7r, size 0603 1 c14 grm31cr60j475ka01l murata (2) 4.7f/6.3v ceramic capacitor, x5r, size 1206 1 d1 bat46w diodes, inc. (4) 100v small signal schottky diode, sod123 1 d2 cmdz5l6 central semi (5) 5.6v zener diode, sod323 1 l1 hcl1305-4r0-r cooper bussmann (6) 4.0h inductor, 10a rms current 1 q1 sir432dp vishay (7) mosfet, n-ch, power so-8 1 q2 sir804dp vishay (7) mosfet, n-ch, power so-8 1 q3 fcx493 zetex (4) 100v npn transistor, sot89 1 r1, r3 crcw060310k0fkea vishay dale (7) 10k ? resistor, size 0603, 1% 2 r2 crcw08051r21fkea vishay dale (7) 1.21? resistor, size 0805, 5% 1 r4 crcw060380k6fkea vishay dale (7) 80.6k ? resistor, size 0603, 1% 1 r5 crcw060340k2fkea vishay dale (7) 40.2k ? resistor, size 0603, 1% 1 r6 crcw060320k0fkea vishay dale (7) 20k ? resistor, size 0603, 1% 1 no tes: 1. epcos: www.epcos.com . 2. murata: www.murata.com . 3. sanyo: www.sanyo.com . 4. diodes inc.: www.diodes.com . 5. central semi: www.centralsemi.com . 6. cooper bussmann: www.cooperbussmann.com . 7. vishay: www.vishay.com .
micrel, inc. mic2176 november 2010 25 m9999-111710-a bill of materials (continued) item part number manufacturer description qty r7 crcw060311k5fkea vishay dale (7) 11.5k ? resistor, size 0603, 1% 1 r8 crcw06038k06fkea vishay dale (7) 8.06k ? resistor, size 0603, 1% 1 r9 crcw06034k75fkea vishay dale (7) 4.75k ? resistor, size 0603, 1% 1 r10 crcw06033k24fkea vishay dale (7) 3.24k ? resistor, size 0603, 1% 1 r11 crcw06031k91fkea vishay dale (7) 1.91k ? resistor, size 0603, 1% 1 r12 crcw060349k24fkea vishay dale (7) 49.9? resistor, size 0603, 1% 1 r13, r21 crcw06030000fkea vishay dale (7) 0 ? resistor, size 0603, 5% 2 r14 crcw08059k7fkea vishay dale (7) 9.7k ? resistor, size 0805, 5% 1 u1 mic2176-2ymm micrel. inc. (8) 75v synchronous buck dc-dc regulator 1 no tes: 8. micrel, inc.: www.micrel.com .
micrel, inc. mic2176 november 2010 26 m9999-111710-a pcb layout figure 8. mic2176 evaluation board top layer
micrel, inc. mic2176 november 2010 27 m9999-111710-a pcb layout (continued) figure 9. mic2176 evaluation bo ard mid-layer 1 (ground plane)
micrel, inc. mic2176 november 2010 28 m9999-111710-a pcb layout (continued) figure 10. mic2176 evaluation board mid-layer 2
micrel, inc. mic2176 november 2010 29 m9999-111710-a pcb layout (continued) figure 11. mic2176 evaluation board bottom layer
micrel, inc. mic2176 november 2010 30 m9999-111710-a recommended land pattern
micrel, inc. mic2176 november 2010 31 m9999-111710-a package information 10-pin msop (mm) micrel, inc. 2180 fortune drive san jose, ca 95131 usa tel +1 (408) 944-0800 fax +1 (408) 474-1000 web http://www.micrel.com micrel makes no representations or warranties with respect to the accuracy or completeness of the information furnished in this data sheet. this information is not intended as a warranty and micrel does not assume responsibility for its use. micrel reserves the right to change circuitry, specifications and descriptions at any time without notice. no license, whether express, implied, arising by estoppel or other wise, to any intellectual property rights is granted by this document. except as provided in micrel?s terms and conditions of sale for such products, mi crel assumes no liability whatsoever, and micrel disclaims any express or implied warranty relating to the sale and/or use of micrel products including l iability or warranties relating to fitness for a particular purpose, merchantability, or infringement of an y patent, copyright or other intellectual p roperty right micrel products are not designed or author ized for use as components in life support app liances, devices or systems where malfu nction of a product reasonably be expected to result in personal injury. life s upport devices or systems are devices or systems that (a) are intended for surgical impla into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significan t injury to the user. a purchaser?s use or sale of micrel products for use in life support appliances, devices or systems is a purchaser?s own risk and purchaser agrees to fully indemnify micrel for any damages resulting from such use or sale. can nt ? 2010 micrel, incorporated.

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